Methods and circuits for asymmetric distribution of channel equalization between devices

ABSTRACT

A transceiver architecture supports high-speed communication over a signal lane that extends between a high-performance integrated circuit (IC) and one or more relatively low-performance ICs employing less sophisticated transmitters and receivers. The architecture compensates for performance asymmetry between ICs communicating over a bidirectional lane by instantiating relatively complex transmit and receive equalization circuitry on the higher-performance side of the lane. Both the transmit and receive equalization filter coefficients in the higher-performance IC may be adaptively updated based upon the signal response at the receiver of the higher-performance IC.

FIELD OF THE INVENTION

The present invention relates generally to the field of communications,and more particularly to high speed electronic signaling within andbetween integrated circuit devices.

BACKGROUND

The performance of many digital systems is limited by theinterconnection bandwidth within and between integrated circuit devices(ICs). High performance communication channels between ICs suffer frommany effects that degrade signals. Primary among them are frequencydependent channel loss (dispersion) and reflections from impedancediscontinuities, both of which lead to inter-symbol interference (ISI).Attempts to address these effects have employed various equalizationschemes at the transmitter and receiver. Ideally, transmit and receiveequalization work together to mitigate the degradation imposed by thechannel, and thus allow increased data rates and/or reduced probabilityof communication errors.

In some systems, memory systems for example, the communicating ICs havean asymmetry to them that complicates optimization of the transmit andreceive equalization schemes applied to counter the effects of thecorresponding channel. For example, a memory controller thatcommunicates with one or more memory devices may benefit from afabrication technology that is different from that best suited formanufacturing the memory devices. It is therefore often the case that amemory controller can employ circuitry that exhibits significantlyhigher performance in speed and power than that of the associated memorydevice or devices. This process performance asymmetry between thecommunicating ICs complicates the task of optimizing equalizationbetween the two types of devices.

BRIEF DESCRIPTION OF THE DRAWINGS

The subject matter disclosed is illustrated by way of example, and notby way of limitation, in the figures of the accompanying drawings and inwhich like reference numerals refer to similar elements and in which:

FIG. 1 illustrates an asymmetrical signaling system 100 in accordancewith one embodiment.

FIGS. 2A and 2B depict hypothetical waveforms illustrative of signals152 and 149 on respective receiver pads Rx1N and Rx2N.

FIG. 3 depicts a high-speed communication system 300 in accordance withanother embodiment.

FIG. 4A depicts equalization control circuitry 340 in accordance withone embodiment.

FIG. 4B details an embodiment of a tap-value generator 425 of FIG. 4Athat generates a tap value using a sign-sign, least-mean-squared (LMS)algorithm.

FIGS. 5A through 5D are hypothetical waveform diagrams used inconnection with FIGS. 3 and 4A to illustrate the process of applyingappropriate receive coefficients RXα[2:1] to DFE 325 to correct for ISIintroduced over channel 125.

FIG. 6 is a flowchart 600 outlining a process by which calibration block410 may calculate pre-cursor transmit coefficient TXα[−1].

FIGS. 7A through 7D are hypothetical waveform diagrams used toillustrate the effects of applying the above-derived transmitcoefficients to signals on transmit channel 120.

FIG. 8 depicts an asymmetrical signaling system 800 in accordance withanother embodiment.

FIG. 9 depicts an asymmetrical signaling system 900 in accordance withyet another embodiment.

FIG. 10 depicts three memory topologies 1000, 1005, and 1010 thatincorporate asymmetrical signaling systems in accordance with variousembodiments.

DETAILED DESCRIPTION

FIG. 1 illustrates an asymmetrical signaling system 100 in accordancewith one embodiment. System 100 includes a first integrated circuit (IC)device 105 coupled to a second IC device 110 via a high-speed signallane 115. In this example, lane 115 includes a transmit channel 120 anda receive channel 125, which may comprise circuit-board traces. Thetransmit and receive channels may be implemented over the same conductoror set of conductors in other embodiments. Device 105 may be fabricatedusing a process that affords considerably higher transmit and receivecircuit density and speed performance than the process employed infabricating device 110. In one embodiment, for example, the minimumfeature size for input/output transistors on device 105 is less than 50%of the minimum feature size for input/output transistors on device 110.In other embodiments, device 105 has at least twice the number ofinterconnect layers as device 110, allowing more compact implementationof complex circuits. Communication system 100 accommodates the resultingperformance asymmetry between devices 105 and 110 by instantiating themajority of the complex equalization circuitry on device 105. Transmitand receive channels 120 and 125 can thus support the same relativelyhigh symbol rate without burdening the second device 110 with the areaand power required to support complex equalization circuits.

Performance asymmetry between communicating devices might be desirablefor reasons other than or in addition to process differences. Forexample, market demands may weigh more heavily on one of a pair ofcommunicating devices, making it economically advantageous to minimizethe complexity of that device. Also, total system cost can be reducedwhere the performance-enhanced circuitry can be instantiated on one or arelative few devices that communicate with a greater number of lesssophisticated devices.

Device 105 includes a transmitter Tx1 that transmits data received froman internal transmit-data input node TxD1 over channel 120 via a firsttransmitter pad Tx1N, and further includes a receiver Rx1 that receivesdata from channel 125 via a first receiver pad Rx1N. Device 110 includesa second transmitter Tx2 that transmits data to receiver Rx1 via asecond transmitter pad Tx2N, channel 125, and first receiver pad Rx1N.Device 110 further includes a second receiver RX2 that receives the datatransmitted from transmitter Tx1 via first transmitter pad Tx1N, channel120, and a second receiver pad Rx2N.

Transmitter Tx1 includes an output driver 141 and a multi-tap transmitequalizer 142. Transmit equalizer 142 is composed of a transmit pipe 143and an array of one or more output drivers or sub-drivers 147 in thedepicted embodiment. Output driver 141 and sub-drivers 147 functioncollectively to drive an equalized version of a transmit data signalTxD1 onto transmitter pad Tx1N, which may be a bond pad of device 105.The particular transmit equalizer in this embodiment may also bereferred to as a finite-impulse-response (FIR) equalizer. Equalizers onthe transmit-side of a communication channel are sometimes referred toas “pre-emphasis equalizers” or “de-emphasis equalizers” because theyemphasize or de-emphasize signal components prior to transmission overthe channel in order to mitigate the degrading effects of the channel.For example, transmit equalization, typically a flattening of the totalamplitude response over a frequency band of interest, can beaccomplished by amplifying (emphasizing) the signal frequency componentsmost sensitive to channel loss, by attenuating (deemphasizing) signalcomponents that are less sensitive to channel loss, or by a combinationof the two. The goal of equalization is typically to reduce or minimizethe effects of ISI observed at the receive side of the channel (e.g. atpad or pads Rx2N). Equalization is typically accomplished by adjusting acharacteristic of a signal in order to mitigate the effects of ISI.

Channels 120 and 125 are assumed to act as simple low-pass filters inthis example. For each transmitted symbol, equalization signals fromsub-drivers 147 combine with the main signal from driver 141 toselectively deemphasize low-frequency signal components relative tohigh-frequency signal components, and thus to compensate in advance forthe low-pass nature of channel 120. Transmitter Tx1 therefore transmitsa signal 148 with a relatively low signal-to-ISI ratio (SIR) on nodeTx1N. Channel 120 filters out much of the ISI, and the resultingpropagated signal arrives at node Rx2N of receiver Rx2 as a relativelyhigh SIR waveform, signal 149. The transmit signal Tx1N conveyed overchannel 120 may be a binary, differential, AC-coupled voltage signal.Other embodiments may employ signals that are e.g. single-ended,multilevel, DC coupled, or current driven. Moreover, the frequencyresponse of a given channel may differ considerably from a simplelow-pass filter. Many variations will be evident to those skilled in theart.

Each of sub-drivers 147 is either a pre-tap sub-driver or post-tapsub-driver. If driver 141 has already transmitted the symbol value atthe input to a given sub-driver, that sub-driver is a post-tapsub-driver; if driver 141 has yet to transmit the symbol value at theinput to the sub-driver, that sub-driver is a pre-tap driver. Transmitpipe 143 might select, for example, N post-tap drivers and one pre-tapdriver; accordingly, the signal at transmitter pad Tx1N would have asignal level according to data values having symbol latencies of −1, 0,1, 2, . . . , N, where the symbol latency of a given data value refersto the number of symbol times by which the data value of the sub-driverlags that through the main driver 141. In an alternate embodiment theentire driver can be realized as a high-speed digital-to-analogconverter (DAC) structure which can drive arbitrarily complex waveforms(to the DAC resolution), and the pre-emphasis may be computed in thedigital domain on signals that are provided to the DAC inputs.

Different numbers of post-tap and pre-tap drivers may be provided inalternative embodiments, thereby allowing for transmit equalizationbased on values having different symbol latencies with respect to themain tap. The respective tap weights of sub-drivers 147 can becontrolled by application of transmit coefficients TxCo to acorresponding port. Each of the transmit coefficients may be adjustedover a range of values to tailor transmitter Tx1 to a particular channeland noise environment. Such adjustments can be applied once, e.g. atpower-up, or can be adapted periodically during system operation toaccount for changes in the system environment that may impactperformance. Methods and circuits for setting transmit coefficients inaccordance with some embodiments are discussed below.

Transmitter Tx2 transmits a signal 150 having a relatively high SIR atnode Tx2N as can be observed by the relatively clean opening of the eyediagram. Channel 125 induces ISI and loss that degrades signal 150, andthus presents a signal 152 having a relatively low SIR on node Rx1N. Anequalizer 154 within receiver Rx1 equalizes signal 152 to compensate forthe channel ISI, and thus produces an equalized signal Veq.Continuous-time equalizers, decision-feedback equalizers (DFE), andpartial response DFE equalizers, among others, are types of equalizersknown to those skilled in the art that may be used for equalizer 154. Adifferential amplifier 156 compares equalized signal Veq with areference voltage Vr, and a sampler 158 periodically samples the outputof amplifier 156 to recover the data RxD1 expressed in signal TxD2 fromtransmitter Tx2.

Receiver Rx1 additionally includes a signal monitor 160 and equalizationcontrol circuitry 162. Signal monitor 160 provides an error signal Errthat can be used as a measure of the signal ISI remaining on equalizedsignal Veq. Equalization control circuitry 162 employs error signal Errand in some embodiments the recovered data RxD1 to adjust receivecoefficients RxCo that control equalizer 154 to minimize the ISI onequalized signal Veq.

Control circuitry 162 further adjusts transmit coefficients TxCo totransmitter Tx1 to minimize the ISI on signal 149, which is at the farend of channel 120. In other words, control circuitry 162 adjuststransmit equalizer 142 such that the waveform conditioning induced onsignal 148 closely compensates for the undesired ISI imposed by channel120, and therefore minimizes the impact of channel-induced ISI at nodeRx2N of device 110 based on the behavior of the receive equalizer inReceiver Rx1.

Transmit coefficients TxCo may be generated based upon thecharacteristics of transmit channel 120. In some such embodiments, asignal monitor on receiver Rx2 and a backchannel communication link maybe used to communicate error signal measurement information from device110 to 105. In the depicted embodiment, control circuitry 162 insteadderives transmit coefficients TxCo from the characteristics of receivechannel 125. Such derivations may be used where the transmit and receivechannels exhibit similar characteristics: e.g. frequency-dependentattenuation, dispersions, and/or reflections. Such assumptions areparticularly valid when channels 120 and 125 are the same channel andtransmission is bi-directional, or when channels 120 and 125 areportions of a parallel bus between device 105 and device 110. Derivingtransmit coefficients from receive-channel characteristics reduces oreliminates the need for evaluating signal quality at receiver Rx2, whichin turn reduces the requisite complexity of second IC device 110.

Turning now to second IC device 110, receiver Rx2 and transmitter Tx2are each part of a transceiver 164 that conveys data to and from somecore logic (not shown), such as one or more blocks of dynamicrandom-access memory cells. Receiver Rx2 may be simple relative toreceiver Rx1, and here includes a preamplifier 170 and a sampler 175.Preamplifier 170 may support data threshold-voltage adjustments andsampler 175 may support timing adjustments, in which case the samplevoltage and timing of receiver Rx2 may be centered within received dataeyes at the input of sampler 175. Though not shown, receiver Rx2 canadditionally include some simple receive-equalization circuitry, thoughthis can be less complex—and consequently less expensive andarea-intensive—than the circuitry of receiver Rx1. Transmitter Tx2 canlikewise provide some transmit equalization, but the circuitry employedfor this purpose can be less complex than the circuitry transmitter Tx1employs for this purpose. For example, transmitter Tx2 can have fewerfilter taps than transmitter Tx1, or have fixed tap settings that areknown to provide a minimal baseline of equalization.

FIGS. 2A and 2B depict hypothetical waveforms illustrative of signals152 and 149 on respective receiver pads Rx1N and Rx2N. Each waveformincludes a respective signal eye that exhibits a signal-to-interferenceratio (SIR) at the sample instant. Given that there are a number of waysto measure the SIR of a signal, the present disclosure refers to theparticular measure of interest here as the “eye SIR,” or just SIR. Eachreference to an SIR pertains to this particular measure.

In FIG. 2A, the voltage levels labeled Lev1 and Lev0 represent theaverage levels representing logic ones and logic zeros, respectively.The standard deviations of the voltage levels in cases where a logic oneor a logic zero is received are denoted by σ1 and σ0, respectively. Theseparation between Lev1 and receiver threshold Vr is the amount ofreceived “signal” for a logic one, and the ratio between this amount andσ1 may be used to compare the signal and interference levels when alogic one is received. Similarly, the ratio of the separation betweenLev0 and Vr to σ0 may be used when a logic zero is received. The lowerof these two ratios is usually a critical factor in the bit error rateof the receiver. Therefore, we define the SIR in equation form:

SIR=Min(|Lev1−Vr|/σ1,|Lev0−Vr|/σ0)  Eq. 1

Transmitter Tx2 is relatively unsophisticated, so the signal at nodeRx1N exhibits considerable noise (e.g. ISI) as a result of thedispersion and reflections of channel 125. As shown, signal 152 atreceiver pad Rx1N exhibits a relatively low SIR. In contrast, therelatively sophisticated equalizer of transmitter Tx1 compensates forthe attenuation of write channel 120 in advance of transmission. Theresulting signal 149 at receiver pad Rx2N therefore exhibits arelatively high SIR as compared with that of signal 152. In oneembodiment, the SIR at receiver pad Rx2N is at least 30% higher than theSIR at receiver pad Rx1N. Other embodiments may adapt equalizers forsimilar objectives, such as maximizing the signal-to-noise difference(vertical eye opening) instead of the signal-to-noise ratio, maximizingthe horizontal eye opening relative to jitter, or minimizing the overallbit-error rate (BER) of the channel.

The symbol rates of signals conveyed over channels 120 and 125 may bemaintained substantially equal despite the performance asymmetry betweendevices 105 and 110. The sophisticated equalization employed bytransmitter Tx1 allows for a high-quality (high SIR) signal at nodeRx2N, and thus facilitates use of a relatively simple receiver Rx2, andthe sophisticated equalization employed by receiver Rx1 allows receiverRx1 to correctly interpret low-quality (low SIR) signals, and thusfacilitates use of a relatively simple transmitter Tx2.

FIG. 3 depicts a high-speed communication system 300 in accordance withanother embodiment. System 300 includes first and second IC devices 305and 310 separated by a signal lane 315. The embodiment of FIG. 3 is insome ways like system 100 of FIG. 1, like-numbered elements being thesame or similar.

First device 305 includes a transmitter 320 that supports transmitequalization, e.g. of the type discussed above in connection withFIG. 1. Device 305 additionally includes a receive equalizer 325, a datasampler 330, a signal monitor 335, and equalization control circuitry340. Second device 310 is similar to second device 110 of FIG. 1: adetailed discussion of device 310 is therefore omitted for brevity.

Transmitter 320 includes a three-tap transmit equalizer, which in turnincludes a FIFO buffer 321 and three coefficient multipliers 322. Thetransmit equalizer is a finite impulse response (FIR) equalizer in thisexample, though other types of equalizers may be used instead of or inaddition to an FIR. Buffer 321 presents pre-tap data TD_(N+1), current(main) data TD_(N), and post-tap data TD_(N−1) to respective coefficientmultipliers 322. Multipliers 322 multiply the outputs of buffer 321 byrespective tap coefficients TXα[1, 0, −1], and the sum of the outputs ofmultipliers 322 is presented on pad or pads Tx1N as the output oftransmitter 320.

Receive equalizer 325 supports two post-cursor filter taps in thisembodiment, though more or fewer may be used in other embodiments.Equalizer 325 is a decision-feedback equalizer (DFE) in this example,though other types of equalizers may be used instead of or in additionto a DFE. For example, equalizer 325 may include one or morepartial-response DFE (PrDFE) taps, for example of the type described inpublished U.S. Patent App. No. 20050111585 to Stojanovic et al. entitled“Partial Response Receiver.” A buffer 345 presents first post-tap dataRD_(N−1) (first post cursor) and second post-tap data (second postcursor) data RD_(N−2) to respective coefficient multipliers 346.Multipliers 346 multiply the outputs of buffer 345 by respective tapcoefficients RXα[2,1], and the sum of the outputs of multipliers 346 issubtracted from the incoming signal at pad or pads RxN1 to produce anequalized signal Veq. Alternatively, receive equalizer 325 could beimplemented as a direct continuous-time, linear equalizer with tunablecoefficients. In general, an equalizer can adjust the SIR at aparticular node in the receiver to achieve some desired characteristic,e.g. a low bit-error rate or, as in the case of a PrDFE, canspeculatively apply multiple ISI corrections and later select theappropriate correction to achieve the desired characteristic.

An amplifier 350 within signal monitor 335 compares signal Veq with aselected data level Dlev, outputting a signal indicative of a logic one(zero) if Veq is greater than (less than) level Dlev. A sampler 355periodically captures the output from amplifier 350 on rising edges of areceive clock signal RClk to produce a series of error samples Err_(N).Error samples Err_(N) are conveyed to equalizer control circuitry 340 asmeasures of signal ISI for equalized signal Veq.

Amplifier 156 compares signal Veq with reference voltage Vr (e.g. zerovolts), outputting a signal indicative of a logic one (zero) if Veq isgreater than (less than) level Vr. Sampler 158 periodically captures theoutput from amplifier 156 on rising edges of receive clock signal RClkto produce a series of data samples Data_(N). Data samples Data_(N) areconveyed to equalizer control circuitry 340 and to any other circuitry(not shown) to which the received data RxD1 is directed. In accordancewith the depicted embodiment, control circuitry 340 employs the data anderror samples to derive data level Dlev, transmit equalizationcoefficients TXα[1,0,−1] for transmitter 320, and receive equalizationcoefficients RXα[2,1] for receive equalizer 325.

FIG. 4A depicts equalization control circuitry 340 in accordance withone embodiment. Control circuitry 340 includes a tap controller 400, adata filter 405, transmit calibration block 410, a transmit-coefficientgenerator 415, and a DAC 417. Tap controller 400 includes a series ofsynchronous storage elements 420 and tap-value generators 425 thattogether generate, from data and error samples Data_(N) and Err_(N), tapcoefficients RXα[2,1,0]. Tap value RXα[0] is a digital measure of theaverage amplitude of the received data symbols Data_(N−1), which DAC 417converts into voltage Dlev. Tap values RXα[2,1] are the receivecoefficients for equalizer 325.

The error comparisons that produce error signals Err_(N) are based uponthe upper signal level defined by voltage Dlev and applied via amplifier350. Tap controller 400 thus only updates the tap values RXα[2,1,0]based upon measurements that take place when the data sample Data_(N−1)is a logic one. Data filter 405 therefore prevents updates to tapcontroller 400 when the sample Data_(N−1) is a logic zero. Otherembodiments can include a second comparator/sampler pair to generateerror samples when Data_(N−1) is a logic zero, such as by comparing theincoming signal Veq with the lower data level −Dlev, or the referencevoltage to comparator 350 can be varied over a number of values orranges of values to facilitate additional testing and error-correctionmethods.

Receive coefficients RXα[2,1,0] are adjusted to account for the transfercharacteristics of receive channel 125. The calibrated receivecoefficients RXα[2,1,0] are therefore measures of the transfer functionfor channel 125. Assuming transmit channel 120 and receive channel 125exhibit similar transfer characteristics, receive coefficientsRXα[2,1,0] are also a measure of the transfer characteristics oftransmit channel 120. Devices in accordance with some embodiments usereceive coefficients to derive suitable transmit coefficients. In theembodiment of FIG. 4A, this derivation additionally takes into account apre-cursor receive coefficient RXα[−1] produced by calibration block 410but not used for receive-side equalization in this example. Coefficientgenerator 415 may then derive transmit coefficients from receivecoefficients e.g. in a manner detailed below.

FIG. 4B details an embodiment of a tap-value generator 425 of FIG. 4Athat generates a tap value using a sign-sign, least-mean-squared (LMS)algorithm. Other algorithms, such as linear or gradient-descent LMS, canbe used in other embodiments. Generator 425 includes an XNOR gate 430, amultiplier 435 that multiplies the output of XNOR gate 430 by a constantμ, an adder 440, and a register 445. XNOR gate 430 compares thecorresponding data and error samples and presents its output tomultiplier 435. The output of XNOR gate 430 represents a logic one fortrue and a logic negative one for false. The data and error samplesrepresent the signs of the sampled values, so XNOR gate 430 has theeffect of multiplying the signs and presenting the resulting product tomultiplier 435. Multiplier 435 multiplies the product from XNOR gate 430by a selected step size μ, which may be tailored for the selected filtertap. Adder 440 adds the output from multiplier 435 to the currentcontents of register 445, which is then updated with the new count.Register 445 thus accumulates a count representative of the alpha valuefor the filter tap associated with the data samples of a particularlatency.

FIGS. 5A through 5D are hypothetical waveform diagrams used inconnection with FIGS. 3 and 4A to illustrate the process of applyingappropriate receive coefficients RXα[2,1] to DFE 325 to correct for ISIintroduced over channel 125. FIG. 5A depicts an idealized transmit pulse500 for which the value expressing the current data sample D_(N) attransmitter pad Tx2N is normalized to a value of one (1.0) and the priorand subsequent data samples D_(N−1) and D_(N+1) are each normalized to avalue of zero (0.0). FIG. 5B depicts, as a waveform 505, a version oftransmit pulse 500 filtered by receive channel 125 and appearing at padRx1N. As compared with pulse 500, pulse 505 is attenuated to a maximumamplitude of 0.5 for the current data sample D_(N), the corruptedversion of which is labeled cD_(N). The pulse is further corrupted bychannel ISI, which leads to erroneous positive signal amplitudes ofapproximately cD_(N+1)=0.12 and cD_(N+2)=0.02 at the two succeedingsymbol times, and cD_(N−1)=0.05 at the preceding symbol time. Theobjective of receive equalization is, in part, to compensate for the ISIeffects at the symbol times succeeding the main symbol time.

FIG. 5C is a waveform diagram 510 in which a receive-coefficientwaveform 515 is shown with the shape of pulse 505 of FIG. 5B toillustrate how the receive coefficients are applied to compensate forISI imposed by channel 125. In the example, the channel 125 imposed ISIcomponents cD_(N+1) and cD_(N+2) of respective amplitudes 0.12 and 0.02at the two symbol times succeeding reception of corrupted data symbolcD_(N). DFE 325 therefore subtracts a coefficient waveform 515 from thereceived pulse 505 to cancel the ISI: DFE 325 subtracts Data_(N)*RXα[1]from the received signal one symbol time after cD_(N) and subtractsData_(N)*RXα[2] from the received signal two symbol times after cD_(N).In this example, RXα[0] is about 0.50, RXα[1] about 0.12, and RXα[2]about 0.02.

FIG. 5D depicts an equalized waveform 520 that is the sum of waveforms505 and 515 of FIG. 5C. Ideally, the compensation provided by DFE 325exactly counteracts the ISI associated with the prior data symbolswithout adversely impacting the current symbol. In practice, however,the application of receive coefficients may impact the current symboleD_(N). Furthermore, ISI associated with the first pre-cursor tap is notcancelled in this example, and therefore leaves a noise artifactcD_(N−1) in waveform 520 one symbol time prior to receipt of the currentsymbol. The two post-tap artifacts are cancelled in this example,however, leaving equalized signal values eD_(N+1) and eD_(N+2) ofamplitude zero.

Returning to FIG. 4A, coefficient generator 415 employs receivecoefficients RXα[1,0,−1] to calculate transmit coefficients TXα[1,0,−1].In the description thus far, however, tap controller 400 has calculatedonly two of the three receive coefficients employed by coefficientgenerator 415. The following discussion shows how equalization controlcircuitry 340 of FIG. 3 can calculate a pre-cursor receive-channelcoefficient RXα[−1] in accordance with one embodiment.

FIG. 6 is a flowchart 600 outlining a process by which calibration block410 may calculate pre-cursor receive-channel coefficient RXα[−1]. First,in step 605, the receive coefficients RXα[2,1,0] are calculated in themanner detailed above. In some embodiments, step 605 is accomplished byfirst holding values RXα[2,1] constant until value RXα[0] reachesequilibrium, at which time voltage Dlev represents a measure of theaverage symbol amplitude for signal Veq. With reference to FIG. 3,voltage Dlev is considered to represent the amplitude of signal Veq whenerror signal Err_(N−1) is equally likely to express a logic one or alogic zero when the corresponding sampled data symbol Data_(N−1)represents a logic one. Once voltage Dlev is established, the other twotap-value generators are enabled to find the remaining receivecoefficients RXα[2,1]. Once calibrated, the values of receivecoefficients RXα[2,1] are held constant (step 607).

Next, in step 610, data filter 405 is set to enable Dlev adjustment whenincoming data expresses the pattern “10” (i.e., symbol Data_(N−1)=1 andsucceeding symbol Data_(N)=0). Per decision 615 and step 620, errorsamples Err_(N−1) are collected and coefficient RXα[0] adjusted untilErr_(N−1) is again 50% 1's and 50% 0's when this pattern is detected.Using the circuitry of FIG. 4A, these adjustments occur automatically astap controller 400 finds the coefficient RXα[0], and consequently thelevel Dlev, specific to “10” data patterns. In step 625, calibrationblock 410 then stores the value of coefficient RXα[0] as RXα10. Theprocess of steps 610 through 625 is repeated for data pattern “11”. Thatis, in step 630 data filter 405 is set to enable Dlev adjustment whenincoming data expresses the pattern “11” (i.e., symbol Data_(N−1)=1 andsucceeding symbol Data_(N)=1). Per decision 635 and step 640, errorsamples Err_(N−1) are collected and coefficient RXα[0], and consequentlylevel Dlev, is adjusted until Err_(N−1) is again 50% 1's and 50% 0's.Calibration block 410 then, in step 645, stores the new value of RXα[0]as RXα11.

With coefficients RXα[2,1] calibrated, the difference between valuesRXα11 and RXα10 can largely be attributed to two times the ISIassociated with the first pre-cursor filter position. Filter coefficientRXα[−1] can therefore be calculated using this difference (step 650). Insome embodiments the difference may be scaled, as by multiplying thedifference by a constant C, or may be otherwise adjusted, for example,to compensate for different transmit characteristics between thecontroller and memory device. Other embodiments employ similartechniques to calculate additional pre- or post-cursor transmit orreceiver filter coefficients. Returning to the hypothetical example ofFIGS. 5A through 5D, it may be seen that corrupted data sample cD_(N−1)has a value of about 0.05, so coefficient RXα[−1] is set to 0.05.

Transmitter 320 of FIG. 3 typically has a maximum output power that isdivided among its three taps. For illustrative purposes, that outputpower may be normalized to a value of one. Coefficient generator 415(FIG. 4A) is therefore constrained, in this embodiment, such that thesum of the absolutes values of the three filter coefficientsTXα[1,0,−1], which represents the total power output of the transmitter,is equal to one. In equation form:

|TXα[1]|+|TXα[0]|+|TXα[−1]|=1  Eq. 2

Assuming similar transmit and receive channels, the followingrelationships between the transmit and receive coefficients can be usedin the embodiment to approximately correct for ISI in channel 120:

TXα[1]/TXα[0]=−RXα[1]/RXα[0]  Eq. 3

TXα[−1]/TXα[0]=−RXα[−1]/RXα[0]  Eq. 4

Combining equations two through four and solving for each transmitcoefficient gives:

TXα[1]=(−RXα[1]/RXα[0])/(1+|RXα[1]/RXα[0]|+|RXα[−1]/RXα[0]|)  Eq. 5

TXα[0]=(1+|RXα[1]/RXα[0]|+|RXα[−1]/RXα[0]|)⁻¹  Eq. 6

TXα[−1]=(−RXα[−1]/RXα[0])/(1+|RXα[1]/RXα[0]|+|RXα[−1]/RXα[0]|)  Eq. 7

The transmit coefficients, including one or more pre-emphasiscoefficients, can thus be derived using receive coefficients as ameasure of transmit-channel characteristics. In the example discussedabove in connection with FIGS. 5A-5D, the hypothetical receivecoefficients RXα[1,0,−1] were about 0.12, 0.5, and 0.05, respectively.Applying these results to equations five through seven givesapproximately: TXα[1]=−0.18, TXα[0]=0.75, and TXα[−1]=−0.07.

A more comprehensive solution if RXα[−2] and RXα[2] are measured (usinga similar method to that in FIG. 6) is to solve the following matrixequation for TXα[1], TXα[0], and TXα[−1]:

$\begin{matrix}{\begin{bmatrix}0 & 1 & 0\end{bmatrix} = {\left\lbrack {{TX}\; {\alpha \left\lbrack {- 1} \right\rbrack}{TX}\; {\alpha \lbrack 0\rbrack}{TX}\; {\alpha \lbrack 1\rbrack}} \right\rbrack \cdot {\quad\begin{bmatrix}{{{RX}\; {\alpha \lbrack 0\rbrack}},{{RX}\; {\alpha \lbrack 1\rbrack}},{{RX}\; {\alpha \lbrack 2\rbrack}}} \\{{{RX}\; {\alpha \left\lbrack {- 1} \right\rbrack}},{{RX}\; {\alpha \lbrack 0\rbrack}},{{RX}\; {\alpha \lbrack 1\rbrack}}} \\{{{RX}\; {\alpha \left\lbrack {- 2} \right\rbrack}},{{RX}\; {\alpha \left\lbrack {- 1} \right\rbrack}},{{RX}\; {\alpha \lbrack 0\rbrack}}}\end{bmatrix}}}} & {{Eq}.\mspace{14mu} 8}\end{matrix}$

This solution must then be normalized to a total amplitude of one bydividing the solution by |TXα[1]|+|TXα[0]|+TXα[−1]|. This techniques canbe used generally for any number of taps on the transmitter equalizer orreceive DFE. For example, the matrix of equation 8 may be extended andsolved for cases where additional ISI components are measured andadditional TX FIR taps are available. In an alternate embodiment, asimple mapping table, which can be realized through a table-lookup ROM,RAM, or through firmware, can provide the means to set TX equalizercoefficients from RX equalizer coefficients.

FIGS. 7A through 7C are hypothetical waveform diagrams used toillustrate the effects of applying the above-derived transmitcoefficients to signals on transmit channel 120. FIG. 7A depicts anidealized transmit pulse 700 for which the value expressing the currentdata sample D_(N) at transmitter pad Tx1N is normalized to a value ofone (1.0) and the respective prior and subsequent data samples D_(N−1)and D_(N+1) are normalized to values of zero (0.0). As noted previously,transmit channel 120 is presumed to be similar to receive channel 125for which receive coefficients RXα[1,0,−1] were derived. This isimportant, as the expected similarity of the transfer characteristicsbetween the receive and transmit channels supports the assumption thattransmit equalizer coefficients can be derived from the receiveequalizer coefficients and/or ISI measurements. Based upon thisassumption, the channel filtered version of pulse 700, as it appears atpad Rx2N, is expected to be relatively similar to pulse 505 of FIG. 5B,and has been illustrated accordingly in FIG. 7B. In other words,assuming the same normalized transmit pulse from transmitters Tx1 andTx2, receiver pad Rx2N will see a pulse response with substantially thesame ISI as would be seen at receiver pad Rx1N. In the example of FIG.5B, the receive channel 125 of lane 115 attenuated symbol D_(N) by half,resulting in a corrupted symbol cD_(N). The receive channel additionallyimposed ISI components cD_(N+2)=0.02, cD_(N+1)=0.12, and cD_(N−1)=0.05,so pulse 705 of FIG. 7B is assumed to have the same ISI components.

FIG. 7C is a waveform diagram 710 in which a pulse waveform 715 isshaped by application of the transmit coefficients calculated byapplying receive coefficients RXα[1,0,−1] to equations two throughseven: TXα[1]=−0.18, TXα[0]=0.75, and TXα[−1]=−0.07. Waveform 715 ispre-emphasized using these tap values to produce a transmit-equalizedpulse in which the combined output power of pre-emphasized valuespD_(N−1), pD_(N), and pD_(N+1) is one. Ideally, the pre-emphasisprovided by the transmit equalizer exactly counteracts the ISI inducedby channel 120 without adversely impacting the current symbol.

FIG. 7D is a waveform diagram 720 depicting a pulse 725 at pad Rx2N.Pulse 725 is the shape of the pulse received at Rx2N when waveform 715of FIG. 7C is transmitted from Tx1N and subsequently filtered by channel120. The current received value of symbol rD_(N) is attenuated to about0.35, while the pre- and post-tap values rD_(N−1) and rD_(N+1) arereduced to approximately zero. Transmitter 320 does not compensate forpost-tap corruption at time N+2, so pulse 725 exhibits a non-zeroartifact rD_(N+2). Convolution of the transmitted waveform in FIG. 7Cwith the response of the channel also results in small non-zeroartifacts rD_(N−2) and rD_(N+3).

FIG. 8 depicts a system 800 in accordance with another embodiment.System 800 includes a first IC 805 coupled to a second IC 810 via abidirectional communication channel 815. IC 805 includes transmitter 820with an M tap FIR equalizer that communicates transmit signals TxD1 to acorresponding receiver 825 on IC 810. Receiver 825 on IC 810 includesfrom zero to R receive DFE taps, where R is less than X. IC 805additionally includes receiver 830 with an X-tap DFE that receivespre-emphasized transmissions of signals TxD2 from a correspondingtransmitter 835 on IC 810. Transmitter 835 on IC 810 includes from zeroto Y transmitter FIR taps, where Y is less than M. In some embodiments,each of M and X is greater than the number of taps in either receiver825 or transmitter 835. Transmit coefficients TxCo can be derived fromcharacteristics of the receive signal, e.g. in the manner describedabove.

The complete channel transfer function from IC 805 to IC 810, whichincludes characteristics of transmitter 820 and receiver 825 (e.g. gain,bandwidth, and edge-rate) may not precisely match the complete channeltransfer function in the opposite direction. This may be true even inthe case where the channel wire is identical in both directions. Thecoefficient mapping between receiver 830 and transmitter 820 can bealtered to accommodate such differences, however. For example, iftransmitter 835 has a known, fixed transfer function, that transferfunction can be de-convolved from the transmit filter settings fortransmitter 820 using e.g. equation 8. Similarly, if receiver equalizer825 has a known transfer function, it too can be de-convolved from thetransmit filter settings. Adjustments can then be made to thecoefficient mapping between receiver 830 and transmitter 820 viatable-lookup or other methods.

FIG. 9 depicts a system 900 in accordance with yet another embodiment.System 900 is similar to system 800 of FIG. 8, like-numbered elementsbeing the same or similar. In system 900, however, IC 805 communicateswith a plurality of IC devices 810 via a multi-drop bus 905. FIG. 9illustrates an additional benefit of embodiments in which relativelycomplex transmit and receive circuitry are instantiated on one end of asignal lane. In the depicted example, one complex transmitter and onecomplex receiver are instantiated on IC 805 so that each instance of IC810 can be implemented using a relatively simpler receiver/transmitterpair. Complexity of one IC can therefore be used to simplify a pluralityof other ICs. As in the example of FIG. 8, transmit coefficients TxCocan be derived from characteristics of the receive signal.

FIG. 10 depicts four memory topologies 1000, 1005, 1010, and 1012 thatincorporate asymmetrical signaling systems in accordance with some ofmany possible embodiments. Topology 1000 includes a memory controller1015, a memory buffer 1020, and a number N of memory ICs 1025[N:1].Memory buffer 1020 includes transmit and receive circuitry 1030 that isrelatively complex as compared with the corresponding transceivercircuitry on memory ICs 1025[N:1]. The system of topology 1000 can thussupport relatively high symbol rates without burdening memory ICs1025[N:1] with the area and power required to support complexequalization circuits.

Topology 1005 is similar to topology 1000, like-numbered elements beingthe same or similar. In topology 1005, a memory buffer 1035 includes twosets of transmit and receive circuitry 1030: one to communicate withmemory controller 1015 and another to communicate with memory ICs1025[N:1]. The system of topology 1000 can thus support relatively highsymbol rates without burdening memory ICs 1025[N:1] or memory controller1015 with the area and power required to support complex equalizationcircuits.

Topology 1010 includes a memory controller 1040, an optional memorybuffer 1045, and a number N of memory ICs 1050[N:1]. Memory controller1040 includes transmit and receive circuitry 1055 that is relativelycomplex as compared with the corresponding transceiver circuitry onbuffer 1045 and/or memory ICs 1050[N:1]. The system of topology 1010 canthus support relatively high symbol rates without burdening buffer 1045and/or memory ICs 1050[N:1] with the area and power required to supportcomplex equalization circuits.

Topology 1012 includes a memory controller 1056, two or more memorybuffers 1060, and a number of memory ICs 1065 for each buffer 1060. Eachmemory buffer 1060 includes transmit and receive circuitry 1030 tocommunicate with memory controller 1055. In other embodiments memorybuffer 1060 may communicate with memory ICs 1065 via transmit andreceive circuitry that is the same or similar to circuitry 1030.

It can be observed that in any of these embodiments which includemultiple memory devices connected to a single TX/RX block that multiplecoefficients may be stored and used for equalization. Each of themultiple coefficients may be tailored for the individual channelsbetween the controller and each memory device. These coefficients can bestored in a memory, a register file or using other means. Thecoefficients may then be selected for use based on the memory devicewith which the controller is communicating.

In the foregoing description and in the accompanying drawings, specificterminology and drawing symbols are set forth to provide a thoroughunderstanding of the present invention. In some instances, theterminology and symbols may imply specific details that are not requiredto practice the invention. For example, the interconnection betweencircuit elements or circuit blocks may be shown or described asmulti-conductor or single conductor signal lines. Each of themulti-conductor signal lines may alternatively be single-conductorsignal lines, and each of the single-conductor signal lines mayalternatively be multi-conductor signal lines. Signals and signalingpaths shown or described may be single-ended or differential, andembodiments of the invention may be adapted for use with binary ormultilevel-pulse-amplitude-modulated (multi-PAM) signals.

An output of a process for designing an integrated circuit, or a portionof an integrated circuit, comprising one or more of the circuitsdescribed herein may be a computer-readable medium such as, for example,a magnetic tape or an optical or magnetic disk. The computer-readablemedium may be encoded with data structures or other informationdescribing circuitry that may be physically instantiated as anintegrated circuit or portion of an integrated circuit. Although variousformats may be used for such encoding, these data structures arecommonly written in Caltech Intermediate Format (CIF), Calma GDS IIStream Format (GDSII), or Electronic Design Interchange Format (EDIF).Those of skill in the art of integrated circuit design can develop suchdata structures from schematic diagrams of the type detailed above andthe corresponding descriptions and encode the data structures oncomputer readable medium. Those of skill in the art of integratedcircuit fabrication can use such encoded data to fabricate integratedcircuits comprising one or more of the circuits described herein.

While the present invention has been described in connection withspecific embodiments, the invention should not be so limited. Forexample,

-   -   1. Receivers in accordance with the invention can incorporate a        variable-gain amplifier to provide DC-offset or automatic-gain        control. Such amplifiers can be configurable or can be        controlled e.g. using Dlev to establish and maintain desired DC        signal levels.    -   2. Transmitters in accordance with the invention can likewise        incorporate a pre-distorted offset to provide compensation for        any receiver DC-offset or automatic-gain control. Such        transmitter pre-distortion can be configurable or can be        controlled e.g. using Dlev mapped and scaled as appropriate for        the transmitter.    -   3. Transmitters and receivers could use test or training        patterns in lieu of live data to calibrate transmit and receive        coefficients.    -   4. Other types of equalizers, such as passive or active        continuous time analog filters or transversal filters, could be        used instead of or in addition to the above-described        equalizers.    -   5. It may be the case that other circumstances drive the desire        to put equalization on one device. Cost, complexity, experience,        and desire to use standard devices on one end could each be        reasons to asymmetrically distribute the equalization burden.        Moreover, some components are shown directly connected to one        another while others are shown connected via intermediate        components. In each instance the method of interconnection, or        “coupling,” establishes some desired electrical communication        between two or more circuit nodes, or terminals. Such coupling        may often be accomplished using a number of circuit        configurations, as will be understood by those of skill in the        art. Therefore, the spirit and scope of the appended claims        should not be limited to the foregoing description. Only those        claims specifically reciting “means for” or “step for” should be        construed in the manner required under the sixth paragraph of 35        U.S.C. §112.

What is claimed is:
 1. A first integrated circuit to communicate with asecond integrated circuit over at least one conductive path, the firstintegrated circuit comprising: a transmitter to transmit informationfrom the first integrated circuit to the second integrated circuit overa first conductive path, the transmitter including a transmit equalizer;a receiver to receive information from the second integrated circuitover one of the first conductive path, or a second conductive path; anda circuit to generate equalization settings for the transmit equalizerresponsive to inter-symbol interference affecting the transmission ofthe information transmitted to the first integrated circuit from thesecond integrated circuit.